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LTC3405 1.5MHz, 300mA Synchronous Step-Down Regulator in ThinSOT
FEATURES
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DESCRIPTIO
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High Efficiency: Up to 96% Very Low Quiescent Current: Only 20A During Operation 300mA Output Current at VIN = 3V 2.5V to 5.5V Input Voltage Range 1.5MHz Constant Frequency Operation No Schottky Diode Required Low Dropout Operation: 100% Duty Cycle 0.8V Reference Allows Low Output Voltages Shutdown Mode Draws < 1A Supply Current 2% Output Voltage Accuracy Current Mode Operation for Excellent Line and Load Transient Response Overtemperature Protected Low Profile (1mm) ThinSOTTM Package
The LTC (R)3405 is a high efficiency monolithic synchronous buck regulator using a constant frequency, current mode architecture. Supply current during operation is only 20A and drops to <1A in shutdown. The 2.5V to 5.5V input voltage range makes the LTC3405 ideally suited for single Li-Ion battery-powered applications. 100% duty cycle provides low dropout operation, extending battery life in portable systems. Switching frequency is internally set at 1.5MHz, allowing the use of small surface mount inductors and capacitors. The internal synchronous switch increases efficiency and eliminates the need for an external Schottky diode. Low output voltages are easily supported with the 0.8V feedback reference voltage. The LTC3405 is available in a low profile (1mm) ThinSOT package. For new designs, refer to the LTC3405A data sheet. For fixed 1.5V and 1.8V output versions, refer to the LTC3405A-1.5/LTC3405A-1.8 data sheet.
APPLICATIO S
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Cellular Telephones Personal Information Appliances Wireless and DSL Modems Digital Still Cameras MP3 Players Portable Instruments
, LTC and LT are registered trademarks of Linear Technology Corporation. ThinSOT is a trademark of Linear Technology Corporation.
TYPICAL APPLICATIO
VIN 2.7V TO 5.5V 4 CIN 2.2F CER
100 95 VIN = 3.6V
LTC3405 1 6 RUN MODE GND 2 VFB 5
22pF
EFFICIENCY (%)
VIN
SW
3
4.7H**
VOUT* 3.3V
90 85 80 75 VIN = 5.5V 70 65 60 0.1 VIN = 4.2V
+
COUT 33F
887k 280k
3405 F01a
*VOUT CONNECTED TO VIN FOR 2.7V < VIN < 3.3V **MURATA LQH3C4R7M34 TAIYO YUDEN LMK212BJ225MG AVX TPSB336K006R0600
Figure 1a. High Efficiency Step-Down Converter
Figure 1b. Efficiency vs Load Current
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1 100 10 OUTPUT CURRENT (mA) 1000
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LTC3405
ABSOLUTE
(Note 1)
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RATI GS
PACKAGE/ORDER I FOR ATIO
TOP VIEW RUN 1 GND 2 SW 3 6 MODE 5 VFB 4 VIN
Input Supply Voltage .................................. - 0.3V to 6V MODE, RUN, VFB Voltages ......................... - 0.3V to VIN SW Voltage .................................. - 0.3V to (VIN + 0.3V) P-Channel Switch Source Current (DC) ............. 400mA N-Channel Switch Sink Current (DC) ................. 400mA Peak SW Sink and Source Current .................... 630mA Operating Temperature Range (Note 2) .. - 40C to 85C Junction Temperature (Note 3) ............................ 125C Storage Temperature Range ................ - 65C to 150C Lead Temperature (Soldering, 10 sec)................. 300C
ORDER PART NUMBER LTC3405ES6 S6 PART MARKING LTXQ
S6 PACKAGE 6-LEAD PLASTIC SOT-23
TJMAX = 125C, JA = 250C/ W
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The q denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25C. VIN = 3.6V unless otherwise specified.
SYMBOL IVFB IPK VFB VOVL VFB VLOADREG VIN IS PARAMETER Feedback Current Peak Inductor Current Regulated Feedback Voltage Output Overvoltage Lockout Reference Voltage Line Regulation Output Voltage Load Regulation Input Voltage Range Input DC Bias Current Pulse Skipping Mode Burst Mode(R) Operation Shutdown Oscillator Frequency RDS(ON) of P-Channel FET RDS(ON) of N-Channel FET SW Leakage RUN Threshold RUN Leakage Current MODE Threshold MODE Leakage Current (Note 5) VFB = 0.7V, Mode = 3.6V, ILOAD = 0A VFB = 0.83V, Mode = 0V, ILOAD = 0A VRUN = 0V, VIN = 5.5V VFB = 0.8V VFB = 0V ISW = 100mA ISW = -100mA VRUN = 0V, VSW = 0V or 5V, VIN = 5V
q q q q q q
CONDITIONS
q
MIN 375
q q q
TYP 500 0.8 50 0.04 0.5
MAX 30 625 0.816 80 0.4 5.5
UNITS nA mA V mV %/V % V A A A MHz kHz A V A V A
VIN = 3V, VFB = 0.7V, Duty Cycle < 35% (Note 4) VOVL = VOVL - VFB VIN = 2.5V to 5.5V (Note 4)
0.784 20
2.5 300 20 0.1 1.2 1.5 210 0.7 0.6 0.01 0.3 0.3 1 0.01 1.5 0.01
400 35 1 1.8 0.85 0.90 1 1.5 1 2 1
fOSC RPFET RNFET ILSW VRUN IRUN VMODE IMODE
Burst Mode is a registered trademark of Linear Technology Corporation. Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC3405E is guaranteed to meet performance specifications from 0C to 70C. Specifications over the -40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: LTC3405: TJ = TA + (PD)(250C/W) Note 4: The LTC3405 is tested in a proprietary test mode that connects VFB to the output of the error amplifier. Note 5: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency.
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LTC3405 TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure1a Except for the Resistive Divider Resistor Values) Efficiency vs Input Voltage
95 90 85
EFFICIENCY (%)
IOUT = 100mA IOUT = 250mA IOUT = 10mA IOUT = 1mA
EFFICIENCY (%)
75 70 65 60 55 50 Burst Mode OPERATION VOUT = 1.8V 2.5 3.0 3.5 4.0 4.5 INPUT VOLTAGE (V) 5.0 5.5 IOUT = 0.1mA
60 50 40 30 20 10 0 0.1
EFFICIENCY (%)
80
Efficiency vs Output Current
100 90 80 70 60 VIN = 4.2V 50 40 0.1 VOUT = 1.3V 1 100 10 OUTPUT CURRENT (mA) 1000
3405 G05
VIN = 2.7V
REFERENCE VOLTAGE (V)
EFFICIENCY (%)
0.804 0.799 0.794 0.789 0.784 -50 -25
FREQUENCY (MHz)
VIN = 3.6V
Oscillator Frequency vs Supply Voltage
1.8 1.834 1.824
OUTPUT VOLTAGE (V)
OSCILLATOR FREQUENCY (MHz)
1.7 1.6 1.5 1.4 1.3 1.2
RDS(0N) ()
2
3 4 5 SUPPLY VOLTAGE (V)
UW
3405 G02
Efficiency vs Output Current
100 90 V = 3.6V IN 80 70 VIN = 3.6V VIN = 4.2V VIN = 4.2V 80 70 60 50 100
Efficiency vs Output Current
VIN = 2.7V 90 VIN = 3.6V VIN = 4.2V VIN = 5.5V
PULSE SKIPPING MODE Burst Mode OPERATION VOUT = 1.8V 1 100 10 OUTPUT CURRENT (mA) 1000
3405 G03
40 0.1
VOUT = 1.8V 1 100 10 OUTPUT CURRENT (mA) 1000
3405 G04
Reference Voltage vs Temperature
0.814 VIN = 3.6V 0.809 1.65 1.60 1.55 1.50 1.45 1.40 1.35 50 25 75 0 TEMPERATURE (C) 100 125 1.70
Oscillator Frequency vs Temperature
VIN = 3.6V
1.30 -50 -25
50 25 75 0 TEMPERATURE (C)
100
125
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Output Voltage vs Load Current
1.2 Burst Mode OPERATION PULSE SKIPPING MODE 1.1 1.0 0.9 1.814 1.804 1.794 1.784 VIN = 3.6V 1.774 6
3405 G08
RDS(ON) vs Input Voltage
0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 500 600 0 0 1
MAIN SWITCH
SYNCHRONOUS SWITCH
0
100
200 300 400 LOAD CURRENT (mA)
3 2 5 4 INPUT VOLTAGE (V)
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7
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LTC3405 TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1a Except for the Resistive Divider Resistor Values) Dynamic Supply Current vs Temperature
600
RDS(ON) vs Temperature
1.2
DYNAMIC SUPPLY CURRENT (A)
DYNAMIC SUPPLY CURRENT (A)
VIN = 4.2V 1.0 V = 2.7V IN 0.8 VIN = 3.6V
RDS(ON) ()
0.6 0.4 0.2 SYNCHRONOUS SWITCH MAIN SWITCH 0 -50 -25 50 25 75 0 TEMPERATURE (C) 100 125
Switch Leakage vs Temperature
160 VIN = 5.5V 140 RUN = 0V
SWITCH LEAKAGE (nA) SWITCH LEAKAGE (pA)
120 100 80 60 40 20 0 -50 -25 SYNCHRONOUS SWITCH MAIN SWITCH
50 25 75 0 TEMPERATURE (C)
Pulse Skipping Mode Operation
SW 5V/DIV
VOUT 20mV/DIV AC COUPLED
IL 100mA/DIV
VIN = 3.6V VOUT = 1.8V ILOAD = 20mA
500ns/DIV
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UW
3405 G11
Dynamic Supply Current
1600 1400 1200 1000 800 600 PULSE SKIPPING MODE 400 200 Burst Mode OPERATION 0 2 3 4 5 SUPPLY VOLTAGE (V) 6
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VOUT = 1.8V ILOAD = 0A
500 400 300 200 100
VIN = 3.6V VOUT = 1.8V ILOAD = 0A PULSE SKIPPING MODE
Burst Mode OPERATION 0 -50 -25 50 25 75 0 TEMPERATURE (C) 100 125
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Switch Leakage vs Input Voltage
60 RUN = 0V 50 40 30 20 MAIN SWITCH 10 0 IL 100mA/DIV SYNCHRONOUS SWITCH SW 5V/DIV
Burst Mode Operation
VOUT 50mV/DIV AC COUPLED
100
125
0
1
2 3 4 INPUT VOLTAGE (V)
5
6
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3405 G14
VIN = 3.6V VOUT = 1.8V ILOAD = 20mA
5s/DIV
3405 G16
Start-Up from Shutdown
VOUT 100mV/DIV AC COUPLED IL 200mA/DIV
Load Step
RUN 2V/DIV
VOUT 1V/DIV
IL 200mA/DIV
ILOAD 200mA/DIV
VIN = 3.6V VOUT = 1.8V ILOAD = 250mA
100s/DIV
3405 G18
VIN = 3.6V 40s/DIV VOUT = 1.8V ILOAD = 0mA TO 250mA PULSE SKIPPING MODE
3405 G19
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LTC3405 TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1a Except for the Resistive Divider Resistor Values) Load Step
VOUT 100mV/DIV AC COUPLED IL 200mA/DIV VOUT 100mV/DIV AC COUPLED IL 200mA/DIV
ILOAD 200mA/DIV
VIN = 3.6V 40s/DIV VOUT = 1.8V ILOAD = 20mA TO 250mA PULSE SKIPPING MODE
PI FU CTIO S
RUN (Pin 1): Run Control Input. Forcing this pin above 1.2V enables the part. Forcing this pin below 0.4V shuts down the device. In shutdown, all functions are disabled drawing <1A supply current. Do not leave RUN floating. GND (Pin 2): Ground Pin. SW (Pin 3): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. VIN (Pin 4): Main Supply Pin. Must be closely decoupled to GND, Pin 2, with a 2.2F or greater ceramic capacitor. VFB (Pin 5): Feedback Pin. Receives the feedback voltage from an external resistive divider across the output. MODE (Pin 6): Mode Select Input. To select pulse skipping mode, tie to VIN. Grounding this pin selects Burst Mode operation. Do not leave this pin floating.
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Load Step
VOUT 100mV/DIV AC COUPLED
Load Step
IL 200mA/DIV
ILOAD 200mA/DIV
ILOAD 200mA/DIV VIN = 3.6V 40s/DIV VOUT = 1.8V ILOAD = 20mA TO 250mA Burst Mode OPERATION VIN = 3.6V 40s/DIV VOUT = 1.8V ILOAD = 0mA TO 250mA Burst Mode OPERATION
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LTC3405
FU CTIO AL DIAGRA
MODE 6 SLOPE COMP OSC OSC
FREQ SHIFT VFB 5 0.8V
EA
VIN RUN 1 0.8V REF 0.85V SHUTDOWN
-
OVDET
+
IRCMP
OPERATIO
(Refer to Functional Diagram)
Main Control Loop The LTC3405 uses a constant frequency, current mode step-down architecture. Both the main (P-channel MOSFET) and synchronous (N-channel MOSFET) switches are internal. During normal operation, the internal top power MOSFET is turned on each cycle when the oscillator sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor current at which ICOMP resets the RS latch, is controlled by the output of error amplifier EA. The VFB pin, described in the Pin Functions section, allows EA to receive an output feedback voltage from an external resistive divider. When the load current increases, it causes a slight decrease in the feedback voltage relative to the 0.8V reference, which in turn, causes the EA amplifier's output voltage to increase until the average inductor current matches the new load current. While the top MOSFET is off, the bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the current reversal comparator IRCMP, or the beginning of the next clock cycle.
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+
-
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0.65V 4 VIN
-
+
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U
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- +
0.4V
- +
EN SLEEP
-
BURST Q Q SWITCHING LOGIC AND BLANKING CIRCUIT
ICOMP
+
5
S R
RS LATCH
ANTISHOOTTHRU
3 SW
OV
2 GND
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LTC3405
OPERATIO
Comparator OVDET guards against transient overshoots > 6.25% by turning the main switch off and keeping it off until the fault is removed. Burst Mode Operation The LTC3405 is capable of Burst Mode operation in which the internal power MOSFETs operate intermittently based on load demand. To enable Burst Mode operation, simply connect the MODE pin to GND. To disable Burst Mode operation and enable PWM pulse skipping mode, connect the MODE pin to VIN or drive it with a logic high (VMODE > 1.5V). In this mode, the efficiency is lower at light loads, but becomes comparable to Burst Mode operation when the output load exceeds 25mA. The advantage of pulse skipping mode is lower output ripple and less interference to audio circuitry. When the converter is in Burst Mode operation, the peak current of the inductor is set to approximately 100mA regardless of the output load. Each burst event can last from a few cycles at light loads to almost continuously cycling with short sleep intervals at moderate loads. In between these burst events, the power MOSFETs and any unneeded circuitry are turned off, reducing the quiescent current to 20A. In this sleep state, the load current is being supplied solely from the output capacitor. As the output voltage droops, the EA amplifier's output rises above the sleep threshold signaling the BURST comparator to trip and turn the top MOSFET on. This process repeats at a rate that is dependent on the load demand. Short-Circuit Protection When the output is shorted to ground, the frequency of the oscillator is reduced to about 210kHz, 1/7 the nominal frequency. This frequency foldback ensures that the inductor current has more time to decay, thereby preventing runaway. The oscillator's frequency will progressively increase to 1.5MHz when VFB rises above 0V. Dropout Operation As the input supply voltage decreases to a value approaching the output voltage, the duty cycle increases toward the maximum on-time. Further reduction of the supply voltage forces the main switch to remain on for more than one cycle
MAXIMUM OUTPUT CURRENT (mA)
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(Refer to Functional Diagram)
until it reaches 100% duty cycle. The output voltage will then be determined by the input voltage minus the voltage drop across the P-channel MOSFET and the inductor. Another important detail to remember is that at low input supply voltages, the RDS(ON) of the P-channel switch increases (see Typical Performance Characteristics). Therefore, the user should calculate the power dissipation when the LTC3405 is used at 100% duty cycle with low input voltage (See Thermal Considerations in the Applications Information section). Low Supply Operation The LTC3405 will operate with input supply voltages as low as 2.5V, but the maximum allowable output current is reduced at this low voltage. Figure 2 shows the reduction in the maximum output current as a function of input voltage for various output voltages.
600 VOUT = 1.8V 500 400 VOUT = 2.5V 300 200 100 0 VOUT = 1.3V
2.5
3.0
3.5 4.0 4.5 SUPPLY VOLTAGE (V)
5.0
5.5
3405 G23
Figure 2. Maximum Output Current vs Input Voltage
Slope Compensation and Inductor Peak Current Slope compensation provides stability in constant frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by adding a compensating ramp to the inductor current signal at duty cycles in excess of 40%. Normally, this results in a reduction of maximum inductor peak current for duty cycles > 40%. However, the LTC3405 uses a patent-pending scheme that counteracts this compensating ramp, which allows the maximum inductor peak current to remain unaffected throughout all duty cycles.
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LTC3405
APPLICATIO S I FOR ATIO
The basic LTC3405 application circuit is shown in Figure 1. External component selection is driven by the load requirement and begins with the selection of L followed by CIN and COUT. Inductor Selection For most applications, the value of the inductor will fall in the range of 3.3H to 10H. Its value is chosen based on the desired ripple current. Large value inductors lower ripple current and small value inductors result in higher ripple currents. Higher VIN or VOUT also increases the ripple current as shown in equation 1. A reasonable starting point for setting ripple current is IL = 120mA (40% of 300mA).
IL =
V 1 VOUT 1 - OUT ( f)(L) VIN
The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 360mA rated inductor should be enough for most applications (300mA + 60mA). For better efficiency, choose a low DC-resistance inductor. The inductor value also has an effect on Burst Mode operation. The transition to low current operation begins when the inductor current peaks fall to approximately 100mA. Lower inductor values (higher IL) will cause this to occur at lower load currents, which can cause a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to increase. Inductor Core Selection Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don't radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3405 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3405 applications.
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Table 1. Representative Surface Mount Inductors
MANUFACTURER PART NUMBER Taiyo Yuden Panasonic Murata Taiyo Yuden Panasonic Panasonic Sumida LB2016T3R3M ELT5KT4R7M LQH3C4R7M34 LB2016T4R7M ELT5KT6R8M ELT5KT100M MAX DC VALUE CURRENT DCR HEIGHT 3.3H 4.7H 4.7H 4.7H 6.8H 10H 280mA 950mA 450mA 210mA 760mA 680mA 620mA 0.2 1.6mm 0.2 1.2mm 0.2 2mm 0.25 1.6mm 0.3 1.2mm 0.36 1.2mm 0.23 1.2mm CMD4D116R8MC 6.8H
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CIN and COUT Selection In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by:
(1)
[VOUT (VIN - VOUT )]1/ 2 CIN required IRMS IOMAX
VIN
This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that the capacitor manufacturer's ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Always consult the manufacturer if there is any question. The selection of COUT is driven by the required effective series resistance (ESR). An ESR in the range of 100m to 200m is necessary to provide a stable loop. For the LTC3405, the general rule for proper operation is: 0.1 COUT required ESR 0.6 ESR is a direct function of the volume of the capacitor; that is, physically larger capacitors have lower ESR. Once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple VOUT is determined by:
1 VOUT IL ESR + 8fC OUT
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LTC3405
APPLICATIO S I FOR ATIO
where f = operating frequency, COUT = output capacitance and IL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since IL increases with input voltage. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. However, care must be taken when these capacitors are used at the input and the output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. When ceramic capacitors are used at the output, their low ESR cannot provide sufficient phase lag cancellation to stabilize the loop. One solution is to use a tantalum capacitor, with its higher ESR, to provide the bulk capacitance and parallel it with a small ceramic capacitor to reduce the ripple voltage as shown in Figure 3.
VIN 2.7V TO 4.2V 4 CIN 2.2F CER VIN RUN MODE GND 2 VFB 5 887k 1M
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SW
3
4.7H 22pF COUT1 + 1F CER
LTC3405 1 6
Figure 3. Paralleling a Ceramic with a Tantalum Capacitor
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Another solution is to connect the feedback resistor to the SW pin as shown in Figure 4. Taking the feedback information at the SW pin removes the phase lag due to the output capacitor resulting in a very stable loop. This configuration lowers the load regulation by the DC resistance of the inductor multiplied by the load current. This slight shift in load regulation actually helps reduce the overshoot and undershoot of the output voltage during a load transient.
VIN 2.7V TO 4.2V 4 CIN 2.2F CER VIN RUN MODE GND 2 VFB 5 SW 3 4.7H 887k 22pF VOUT 1.5V COUT 4.7F CER LTC3405 1 6 1M
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Figure 4. Using All Ceramic Capacitors
A third solution is to use a high value resistor to inject a feedforward signal at VFB mimicking the ripple voltage of a high ESR output capacitor. The circuit in Figure 5 shows how this technique can be easily realized. The feedforward resistor, R2B, is connected to SW as in the previous example. However, in this case, the feedback information is taken from the resistive divider, R2A and R1, at the output. This eliminates most of the load regulation degradation due to the DC resistance of the inductor while providing a stable operation similar to that obtained from a high ESR tantalum type capacitor. Using this technique, the extra feedforward resistor, R2B, must be accounted for when calculating the resistive divider as follows:
R2 = R2A || R2B = VOUT
VIN 2.7V TO 4.2V
R2A * R2B R2A + R2B R2 = 0.8V 1 + R1
4 CIN 2.2F CER VIN RUN MODE GND 2 VFB 5 R2A R1 215k 200k
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VOUT 1.5V COUT2 22F TANT
SW
3
4.7H R2B 22pF 1M
LTC3405 1 6
VOUT 1.5V COUT1 4.7F CER
Figure 5. Feedforward Injection in an All Ceramic Capacitor Application
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LTC3405
APPLICATIO S I FOR ATIO
In pulse skipping mode, the LTC3405 is stable with a 4.7F ceramic output capacitor with VIN 4.2V. For single Li-Ion applications operating in pulse skipping mode, the circuit shown in Figure 6 can be used When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size.
VIN 2.7V TO 4.2V 4 CIN 2.2F CER VIN RUN MODE GND 2 VFB 5 887k 1M
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SW
3
4.7H 22pF COUT1 4.7F CER
LTC3405 1 6
POWER LOST (W)
Figure 6. Using All Ceramic Capacitors in Pulse Skipping Mode
Output Voltage Programming The output voltage is set by a resistive divider according to the following formula: VOUT R2 = 0.8V 1 + R1
(2)
0.0001 0.1
The external resistive divider is connected to the output, allowing remote voltage sensing as shown in Figure 7.
0.8V VOUT 5.5V R2 VFB LTC3405 GND
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R1
Figure 7. Setting the LTC3405 Output Voltage
Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% - (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power.
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Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses in LTC3405 circuits: VIN quiescent current and I2R losses. The VIN quiescent current loss dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power lost is of no consequence as illustrated in Figure 8.
1 VIN = 3.6V
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VOUT 1.5V
0.1 VOUT = 1.8V 0.01
0.001
VOUT = 2.5V
VOUT = 3.3V
VOUT = 1.3V 1 100 10 LOAD CURRENT (mA) 1000
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Figure 8. Power Lost vs Load Current
1. The VIN quiescent current is due to two components: the DC bias current as given in the electrical characteristics and the internal main switch and synchronous switch gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. In continuous mode, IGATECHG = f(QT + QB) where QT and QB are the gate charges of the internal top and bottom switches. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. In continuous mode, the average output current flowing through inductor L is "chopped" between the main switch and the synchronous switch. Thus, the series resistance looking into the SW pin is a function of both
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LTC3405
APPLICATIO S I FOR ATIO
top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 - DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Charateristics curves. Thus, to obtain I2R losses, simply add RSW to RL and multiply the result by the square of the average output current. Other losses including CIN and COUT ESR dissipative losses and inductor core losses generally account for less than 2% total additional loss. Thermal Considerations In most applications the LTC3405 does not dissipate much heat due to its high efficiency. But, in applications where the LTC3405 is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150C, both power switches will be turned off and the SW node will become high impedance. To avoid the LTC3405 from exceeding the maximum junction temperature, the user will need to do a thermal analysis. The goal of the thermal analysis is to determine whether the operating conditions exceed the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(JA) where PD is the power dissipated by the regulator and JA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: TJ = TA + TR where TA is the ambient temperature. As an example, consider the LTC3405 in dropout at an input voltage of 2.7V, a load current of 300mA and an
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ambient temperature of 70C. From the typical performance graph of switch resistance, the RDS(ON) of the P-channel switch at 70C is approximately 0.94. Therefore, power dissipated by the part is: PD = ILOAD2 * RDS(ON) = 84.6mW For the SOT-23 package, the JA is 250C/ W. Thus, the junction temperature of the regulator is: TJ = 70C + (0.0846)(250) = 91.15C which is well below the maximum junction temperature of 125C. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)). Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to (ILOAD * ESR), where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT, which generates a feedback error signal. The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability problem. For a detailed explanation of switching control loop theory, see Application Note 76. A second, more severe transient is caused by switching in loads with large (>1F) supply bypass capacitors. The discharged bypass capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the load switch resistance is low and it is driven quickly. The only solution is to limit the rise time of the switch drive so that the load rise time is limited to approximately (25 * CLOAD). Thus, a 10F capacitor charging to 3.3V would require a 250s rise time, limiting the charging current to about 130mA.
3405f
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LTC3405
APPLICATIO S I FOR ATIO
PC Board Layout Checklist
When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3405. These items are also illustrated graphically in Figures 9 and 10. Check the following in your layout: 1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept short, direct and wide. 2. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground. 3. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs.
1
-
VOUT COUT
+
L1
BOLD LINES INDICATE HIGH CURRENT PATHS *ADD R3 FOR APPLICATIONS USING A CERAMIC COUT
Figure 9. LTC3405 Layout Diagram
VOUT
PIN 1 L1 LTC3405 SW
*ADD R3 WHEN USING CERAMIC COUT
Figure 10. LTC3405 Suggested Layout
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4. Keep the switching node, SW, away from the sensitive VFB node. Design Example As a design example, assume the LTC3405 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to about 2.7V. The load current requirement is a maximum of 0.25A but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low and high load currents is important. Output voltage is 2.5V. With this information we can calculate L using equation (1),
L= V 1 VOUT 1 - OUT ( f)(IL ) VIN
(3)
RUN MODE 6 LTC3405 2 GND VFB VIN CIN R3* 5 R2 3 SW 4 CFWD R1
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+
VIN
-
3405 F09
VIA TO SW NODE
VIA TO GND
R3* VIA TO VIN
VFB
R1 VIN VIA TO VOUT
R2 CFWD
COUT GND
CIN
3405 F10
LTC3405
APPLICATIO S I FOR ATIO
Substituting VOUT = 2.5V, VIN = 4.2V, IL = 100mA and f = 1.5MHz in equation (3) gives: L= 2.5V 2.5V 1 - 6.8H 1.5MHz(100mA) 4.2V
For best efficiency choose a 300mA or greater inductor with less than 0.3 series resistance. CIN will require an RMS current rating of at least 0.125A ILOAD(MAX)/2 at temperature and COUT will require an ESR of less than 0.6 and greater than 0.1. In most cases, a tantalum capacitor will satisfy this requirement.
VIN 2.7V TO 4.2V 4 CIN 2.2F CER
EFFICIENCY (%)
VIN RUN MODE GND 2
SW
3
6.8H* 22pF
VOUT 2.5V
LTC3405 1 6 5
+
COUT** 33F TANT
VFB
887k 412k
3405 F11a
*SUMIDA CMD4D11-6R8MC ** AVX TPSB336K006R0600 TAIYO YUDEN LMK212BJ225MG
Figure 11a
TYPICAL APPLICATIO S
Single Li-Ion to 1.8V/300mA Regulator Optimized for Small Footprint and High Efficiency
VIN 2.7V TO 4.2V 4 CIN** 1F CER VIN RUN MODE GND 2 VFB 5 SW 3 1M 4.7H* 22pF VOUT 1.8V COUT 4.7F CER
1 6
100 90 80 EFFICIENCY (%) 70 60 50 40 30 0.1 VIN = 4.2V VIN = 2.7V VIN = 3.6V
1 100 10 OUTPUT CURRENT (mA)
1000
3405 TA01b
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For the feedback resistors, choose R1 = 412k. R2 can then be calculated from equation (2) to be: V R2 = OUT - 1 R1 = 875.5k; use 887k 0.8 Figure 11 shows the complete circuit along with its efficiency curve.
100 VIN = 2.7V 90 VIN = 3.6V 80 70 60 50 40 30 0.1 VIN = 4.2V 1 100 10 OUTPUT CURRENT (mA) 1000
3405 F11b
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Figure 11b
LTC3405
200k
332k *MURATA LQH3C4R7M34 **TAIYO YUDEN CERAMIC JMK107BJ105MA 3405 TA01a TAIYO YUDEN CERAMIC JMK212BJ475MG
VOUT 100mV/DIV AC COUPLED
IL 200mA/DIV
ILOAD 200mA/DIV
VIN = 3.6V 40s/DIV ILOAD = 100mA TO 250mA
3405 TA01c
3405f
13
LTC3405
TYPICAL APPLICATIO S
Single Li-Ion to 1.8V/300mA Regulator Using Ceramic and Tantalum Output Capacitors
VIN 2.7V TO 4.2V 4 CIN** 2.2F CER VIN RUN MODE GND 2 VFB 5 887k SW 3 4.7H* 22pF COUT1*** + 1F CER VOUT 1.8V COUT2 22F TANT
100 VIN = 2.7V 90 80 EFFICIENCY (%) 70 60 50 40 30 0.1 VIN = 4.2V VIN = 3.6V
VOUT 100mV/DIV AC COUPLED
1 100 10 OUTPUT CURRENT (mA)
Single Li-Ion to 1.8V/200mA Regulator Using All Ceramic Capacitors Optimized for Smallest Footprint
VIN 2.7V TO 4.2V 4 CIN** 1F CER VIN RUN MODE GND 2 VFB 5 SW 3 1M 3.3H* 22pF VOUT 1.8V COUT 4.7F CER
100 90 80 EFFICIENCY (%) VIN = 3.6V 70 60 50 40 30 0.1 VIN = 4.2V
ILOAD 200mA/DIV IL 200mA/DIV
VIN = 2.7V
1 100 10 OUTPUT CURRENT (mA)
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LTC3405 1 6
698k
*MURATA LQH3C4R7M34 **TAIYO YUDEN CERAMIC LMK212BJ225MG ***TAIYO YUDEN CERAMIC JMK107BJ105MA AVX TAJA226M006R 3405 TA02a
IL 200mA/DIV
ILOAD 200mA/DIV
1000
3405 TA02b
VIN = 3.6V 40s/DIV VOUT = 1.8V ILOAD = 100mA TO 250mA
3405 TA02c
LTC3405 1 6
332k 200k *TAIYO YUDEN LB2016T3R3M **TAIYO YUDEN CERAMIC JMK107BJ105MA 3405 TA03a TAIYO YUDEN CERAMIC JMK212BJ475MG
VOUT 100mV/DIV AC COUPLED
1000
3405 TA03b
VIN = 3.6V 40s/DIV ILOAD = 100mA TO 250mA
3405 TA03c
3405f
LTC3405
TYPICAL APPLICATIO S
Single Li-Ion to 1.8V/300mA Regulator Using All Ceramic Capacitors Optimized for Lowest Profile, 1.2mm High
VIN 2.7V TO 4.2V 4 CIN** 1F CER VIN RUN MODE GND 2 VFB 5 SW 3 1M 4.7H* 22pF COUT** 1F CER VOUT 1.8V COUT** 1F CER
100 90 80
EFFICIENCY (%)
VIN = 2.7V
70 60 50 40 30 0.1
VIN = 3.6V VIN = 4.2V
1 100 10 OUTPUT CURRENT (mA)
PACKAGE DESCRIPTIO
SOT-23 (Original) A A1 A2 L .90 - 1.45 (.035 - .057) .00 - 0.15 (.00 - .006) .90 - 1.30 (.035 - .051) .35 - .55 (.014 - .021)
SOT-23 (ThinSOT) 1.00 MAX (.039 MAX) .01 - .10 (.0004 - .004) .80 - .90 (.031 - .035) .30 - .50 REF (.012 - .019 REF) 2.60 - 3.00 (.102 - .118) 1.50 - 1.75 (.059 - .069) (NOTE 3) PIN ONE ID
.20 (.008) DATUM `A' A A2
L NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES)
3. DRAWING NOT TO SCALE 4. DIMENSIONS ARE INCLUSIVE OF PLATING 5. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 6. MOLD FLASH SHALL NOT EXCEED .254mm 7. PACKAGE EIAJ REFERENCE IS: SC-74A (EIAJ) FOR ORIGINAL JEDEC MO-193 FOR THIN
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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LTC3405 1 6
200k
332k *PANASONIC ELT5KT4R7M **TAIYO YUDEN CERAMIC JMK107BJ105MA
3405 TA04a
VOUT 100mV/DIV AC COUPLED
IL 200mA/DIV
ILOAD 200mA/DIV
1000
3405 TA04b
VIN = 3.6V 40s/DIV ILOAD = 100mA TO 250mA
3405 TA04c
S6 Package 6-Lead Plastic SOT-23
(Reference LTC DWG # 05-08-1634) (Reference LTC DWG # 05-08-1636)
2.80 - 3.10 (.110 - .118) (NOTE 3)
.95 (.037) REF
.25 - .50 (.010 - .020) (6PLCS, NOTE 2)
.09 - .20 (.004 - .008) (NOTE 2)
1.90 (.074) REF
A1
S6 SOT-23 0401
3405f
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LTC3405
TYPICAL APPLICATIO S
Single Li-Ion to 1.8V/300mA Regulator All Ceramic Capacitors with Lowest Parts Count
VIN 2.7V TO 4.2V 4 CIN** 2.2F CER 3 4.7H* 887k 22pF VFB 2 5 VOUT 1.8V COUT 4.7F CER
100 VIN = 2.7V 90 VIN = 4.2V 80
EFFICIENCY (%)
70 60 50 40 30 0.1
VIN = 3.6V
1 100 10 OUTPUT CURRENT (mA)
RELATED PARTS
PART NUMBER LTC1174/LTC1174-3.3 LTC1174-5 LTC1265 LTC1474/LTC1475 LTC1504A LT1616 LTC1627 LTC1701 LTC1707 LTC1767 LTC1779 LTC1877 LTC1878 LTC3404 LTC3405A LTC3405A-1.5/ LTC3405A-1.8 DESCRIPTION High Efficiency Step-Down and Inverting DC/DC Converters 1.2A, High Efficiency Step-Down DC/DC Converter Low Quiescent Current Step-Down DC/DC Converters Monolithic Synchronous Step-Down Switching Regulator 600mA, 1.4MHz Step-Down DC/DC Converter Monolithic Synchronous Step-Down Switching Regulator Monolithic Current Mode Step-Down Switching Regulator Monolithic Synchronous Step-Down Switching Regulator 1.5A, 1.25MHz Step-Down Switching Regulator Monolithic Current Mode Step-Down Switching Regulator High Efficiency Monolithic Step-Down Regulator High Efficiency Monolithic Step-Down Regulator 1.4MHz High Efficiency Monolithic Step-Down Regulator 1.5MHz High Efficiency Monolithic Step-Down Regulator 1.5MHz High Efficiency Monolithic Step-Down Regulator COMMENTS Monolithic Switching Regulators, I OUT to 450mA, Burst Mode Operation Constant Off-Time, Monolithic, Burst Mode Operation Monolithic, IOUT to 250mA, IQ = 10A, 8-Pin MSOP Low Cost, Voltage Mode IOUT to 500mA, VIN from 4V to 10V 6-Pin ThinSOT, VIN from 3.6V to 25V Constant Frequency, IOUT to 500mA, Secondary Winding Regulation, VIN from 2.65V to 8.5V Constant Off-Time, IOUT to 500mA, 1MHz Operation, VIN from 2.5V to 5.5V 1.19V VREF Pin, Constant Frequency, IOUT to 600mA, VIN from 2.65V to 8.5V 3V to 25V Input, 8-Lead MSOP Package 550kHz, 6-Lead ThinSOT, V IN from 2.5V to 9.8V 550kHz, MS8, VIN Up to 10V, IQ = 10A, IOUT to 600mA at VIN = 5V 550kHz, MS8, VIN Up to 6V, IQ = 10A, IOUT to 600mA at VIN = 3.3V 1.4MHz, MS8, VIN Up to 6V, IQ = 10A, IOUT to 600mA at VIN = 3.3V Stable with Ceramic Output Capacitor Fixed Output Version of LTC3405A
16
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 q FAX: (408) 434-0507
q
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VIN RUN MODE GND
SW
LTC3405 1 6
698k
3405 TA05a
*MURATA LQH3C4R7M34 **TAIYO YUDEN CERAMIC LMK212BJ225MG TAIYO YUDEN CERAMIC JMK212BJ475MG
VOUT 100mV/DIV AC COUPLED
IL 200mA/DIV
ILOAD 200mA/DIV
1000
3405 TA05b
VIN = 3.6V 40s/DIV ILOAD = 100mA TO 250mA
3405 TA05c
3405f LT/TP 0302 2K * PRINTED IN USA
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2001

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